1. Field
The present disclosure is generally related to the field of transmitters for wireless communication, whereby the transmitter is provided with harmonic-rejection means.
2. Description of the Related Technology
Transmitters operable according to some modern wireless communication standards are based on the well-known principles of direct up-conversion as presented in the scheme of FIG. 1. A baseband signal, in FIG. 1 represented with its in-phase and quadrature components I 110 and Q 112, is multiplied with a suitable local oscillator (LO) 140 signal, summed and then applied to a power amplifier (PA) 170.
Due to the increasing demand for communication bandwidth combined with the scarceness of free spectrum, the complexity and versatility of fourth generation (4G) modulation schemes is greater than ever. One type of 4G wireless service is Long Term Evolution (LTE), which is a standard for wireless communication of high-speed data for mobile phones and data terminals. The goal of LTE is to increase the capacity and speed of wireless data networks.
LTE uses Single Carrier Frequency Division Multiple Access (SC-FDMA) based on Orthogonal Frequency Division Multiple Access (OFDMA) technology as the uplink transmission scheme in order to reduce the peak-to-average power ratio (PAPR) of the transmitted signal. In particular, the LTE standard defines multiple RF bands and groups OFDM modulated subcarriers into Resource Blocks (RB) 216 which can be flexibly used within the allocated channel bandwidth 204. A Resource Block 216 consists of 12 OFDM sub-carriers and occupies 180 kHz bandwidth (BW). Multiple resource blocks can be combined contiguously 240 for the uplink (FIG. 2). When the transmit power is concentrated in a single or in a few RB located at a frequency fBB away from LO frequency fLO, the transmitter performance may be affected. Counter-intermodulation products (C-IM) originating from the limited linearity of the transmitter's baseband (BB) may fall directly or through cross-modulation due to the non-linearity of the power amplifier into the receive band and degrade the frequency division duplexing (FDD) performance. They may also fall into protected bands and violate spectral emission requirements. More in particular, due to 3rd order non-linearity in the transmit path, a counter-IM3 (C-IM3) product appears at the output of the RF circuit at −3*fBB from the LO frequency.
The desired signal and this C-IM3 signal generate intermodulation products in the high power amplifier (PA) 380 following the transmitter. These PA-folded C-IM3 components appear at +4fBB from the wanted signal and −4fBB away from the C-IM3 signal (FIG. 3A). Again these C-IM3 generated products may degrade the FDD performances when disturbing the RX band and/or increase out-of-band spurious emissions, and even violate spectral emission requirements.
Several dominating causes can be indicated for the generation of C-IM3 components. A first and main contribution comes from the baseband 3rd order non-linearity. A second contribution is caused by the intermodulation between the wanted signal at RF and the up-converted baseband at the LO's 3rd order harmonic.
Concerning the first cause, the following is to be noted. A baseband signal with frequency fBB applied to a non-linear baseband section generates third order harmonic distortion (HD) components (BB HD3). This happens both in the I and in the Q path. As the Q baseband signal has a phase difference of 90 degrees with the I signal, the BB HD3 component of the quadrature signal Q has a phase difference of 270 degrees with the BB HD3 of the in-phase signal I. As illustrated in FIG. 3A, the baseband's harmonic (BB HD3) is up-converted and combined in the mixers. In a typical transmit chain setup, signal swings are kept rather high to maintain a good SNR and in combination with low power supply voltages in nm CMOS it is difficult to obtain an extremely linear baseband circuit. The resulting generation of harmonic components at fLO−3fBB as described above turns out to be a critical issue.
A less important contribution originates in the mixer 350, where up-conversion of the baseband (fBB) with the 3rd harmonic of the LO signal (3*fLO), which is due to the mixing with a square wave, creates a component at 3*fLO−fBB, that can generate C-IM3 due to intermodulation with the wanted signal at fLO+fBB in the non-linear PA 370 (FIG. 3B). Indeed, one of the intermodulation products of (3*fLO−fBB) and (fLO+fBB) is located at (3*fLO−fBB)−2*(fLO+fBB), which is exactly fLO−3*fBB., i.e. the same frequency of C-IM3. This is especially true in passive mixers as extensively used in modern modulators, as in these mixers the LO signal is mostly a square wave, which features a significant 3rd harmonic.
The C-IM3 described above may further generate components at fLO+5*fBB and fLO−7*fBB through intermodulation with the wanted signal in the subsequent High Power Amplifier (HPA) 380 (FIG. 3C).
These C-IM3 products should be attenuated as much as possible as they degrade the transceiver performance. C-IM3 performance has only recently been recognized as important and only few publications deal with the problem. However, solving the issue in an efficient way gives a clear advantage. A conventional and obvious way to improve C-IM3 performance is by increasing the baseband's intrinsic linearity. However, this increases the design effort and power consumption, which is obviously detrimental for the on-time between battery reloads. Baseband predistortion to compensate the non-linearity is not appropriate, because of the required wider filter bandwidths and the associated penalty in out-of-band noise. An example of compensation by predistortion can be found in U.S. Pat. No. 6,731,693.
As discussed above, the upconversion of the baseband signal with the third order harmonic of the LO, which is due to the mixing with a square wave, is one of the contributors to the C-IM3 problem. There are two well-known techniques to reduce this HD3 component, but they both suffer from the major drawback that they actually only counter the HD3 generated C-IM3 component and do not improve the baseband generated one.
The first is to trap the HD3 by using an LC-notch filter behind the mixers. The introduction of an LC notch filter results in a significant area increase and, even more importantly, a limitation of the RF frequency range in which the transmitter can operate, which is an important disadvantage, especially for 3GPP standards, which cover multiple RF frequency bands. Furthermore, the baseband linearity still needs to be sufficient to avoid direct C-IM3 generation (due to BB HD3).
The other technique consists in the use of harmonic rejection mixers. An example of this approach is given in the paper “A 1.75-GHz Highly Integrated Narrow-Band CMOS Transmitter With Harmonic-Rejection Mixers” (J. A. Weldon et al., IEEE J. Solid-State Circuits, vol. 36, pp. 2003-2015, Dec. 2001), where harmonic rejection mixers are applied at RF frequencies to attenuate the third and fifth order local oscillator harmonics. The principle is illustrated in FIG. 4, where in the mixers 450, 452, 454 (having) the baseband signal is up-converted to RF frequency by means of LO signals with the same frequency but different phase. Thus, the wanted signal is combined from three phases RF1, RF2 and RF3. While the various phases of the wanted signal are combined constructively, the phases of the 3rd harmonics of the corresponding LO are such that, when multiplied with the ratio 1−√2−1 for the various phases, respectively, they cancel each other, and the LO HD3 component is removed. As a result, the baseband is not upconverted to 3*fLO (and thus the (3*fLO−fBB) product is not generated), and the related C-IM3 mechanism is cancelled. However, this scheme has no impact on the C-IM3 baseband non-linearity related generation and consequently a C-IM3 product due to the presence of BB HD3 component is not eliminated.
Application US2010/255868 presents a solution for controlling the uplink transmit power. In this way high power counter-IM3 signals are avoided. However, the proposed approach rather aims at avoiding the problem by modifying the transmission scheme and RB allocation rather than to improve the transmitter (TX) to allow these transmissions.
In US2011/143697 In-Phase (I) and Quadrature (Q) signals passing from a modem into a direct conversion transmitter are pre-distorted separately from, and independently of, one another. The I signal is pre-distorted to compensate for nonlinearities in the baseband I path circuitry between the modem and the upconverter. The Q signal is pre-distorted to compensate for nonlinearities in the baseband Q path circuitry between the modem and the upconverter. This is similar to traditional predistortion with the corresponding disadvantages described previously. Wide bandwidth in filters results in more power and higher out-of-band noise.
Hence, there is a need for a solution where the C-IM3 generated from the baseband component is reduced so that sensitivity degradation at the receiver is avoided in a FDD scheme and spectral emission masks (out-of-band spectral emission requirements) are respected.